This application guide describes the basic principle of low frequency network analysis through the introduction of network analyzer. Here we mainly introduce the simple measurement of low-frequency 2-port devices, high impedance detection technology and large attenuation measurement.

Basic measurement configuration of 50 Ω DUT

Firstly, for the configuration of using low-frequency network analyzer to measure the transmission characteristics of 2-port devices, we briefly introduce the connection method of typical devices under test. The first case is to measure the transmission response characteristics of 50 Ω devices, such as filters and cables. Figure 2 shows the configuration of such tests using the gain phase test port of the instrument. The r-channel receiver (VR) is used to measure the output voltage of the excitation source at the 50 Ω system impedance (the voltage of the input signal of the 50 Ω transmission line), and the t-channel receiver (VT) is used to measure the voltage of the output signal after transmission through the tested device, and then the instrument calculates the measured voltage ratio (VT / VR) to obtain the transmission coefficient S21.

Figure 3 shows the configuration of measurement with the S-parameter test port of the instrument. There are multiple built-in directional bridges behind the S-parameter test port, so there is no need to use the power separator in the external access measurement configuration in Figure 2. In most cases, the S-parameter test port is used to measure the transmission response characteristics of 50 Ω devices. Shide Technology: capacitance measurement principle - test parameters Chapter 8} Figure 3 is the configuration of measuring with the S parameter test port of the instrument. There are multiple built-in directional bridges behind the S-parameter test port, so there is no need to use the power separator in the external access measurement configuration in Figure 2. In most cases, the S-parameter test port is used to measure the transmission response characteristics of 50 Ω devices.

For the test of transmission response characteristics of most 50 Ω devices, use the S parameter test port of the instrument. However, for the measurement of large attenuation devices, for example, when measuring the impedance of DC - DC converter and large capacitance bypass capacitor with only milliohm level, it is usually necessary to adopt the measurement method of shunt thru. This measurement of transmission response characteristics needs to be measured by using the beneficial phase test port of the instrument rather than the S-parameter test port. In this case, the semi floating ground structure of the gain phase test port receiver can avoid the measurement error in the low frequency range, which is caused by the grounding loop of the test cable between the excitation signal source and the receiver (described in detail later).

Fig. 2 measurement configuration for measuring transmission coefficient of 50 Ω device under test using gain phase test port

Fig. 3 measurement configuration for measuring transmission coefficient of 50 Ω device under test using S-parameter test port

Basic measurement configuration

Non 50 Ω DUT, example 1

Low frequency 2-port devices usually have a non-50 Ω impedance, and the low-frequency amplifier circuit is a typical example. Fig. 4 is a measurement configuration example for measuring the frequency response characteristics of a low frequency amplifier using a gain phase test port. The input impedance of the device under test is very high, and the output port is connected to a non-50 Ω load ZL. According to the requirements of practical application, the load impedance ZL can be resistive load or reactance load.

The parameter to be measured is the voltage transfer function from the input port to the output port of the device under test, i.e. / out / / in. Different from the measurement of the transmission coefficient of the 50 Ω device shown in Fig. 2 and Fig. 3, the r-channel receiver (VR) directly measures the AC voltage on the input impedance Zin of the tested device using a high impedance detection method, rather than measuring the voltage on the 50 Ω system impedance. Using high impedance detection, the output voltage (VOUT) can be measured without affecting the load of the device under test.

According to the required maximum measurement frequency, the input impedance of the probe, the input capacitance of the probe, etc. (which will be introduced later), the high impedance measurement receiver of the instrument can be connected with the tested device with a coaxial measurement cable or a 10:1 passive probe. When using coaxial test cable, a T-connector can be used at the r-channel detection point. In order to compensate the frequency response and phase error between two probes / test cables, it is necessary to calibrate the through response by placing the probe point connected with T channel on the TPI test point, and then measuring.

Figure 4 configuration of measurement amplifier using gain phase port (maximum measurement frequency up to 30 MHz)

If you want to measure the frequency response of the amplifier at a measurement frequency above 30 MHz, or you need to use a probe with very small capacitance to measure the amplifier, use an active probe to measure the S parameter test port of the instrument, as shown in Figure 5. Different from the configuration in Fig. 4, the ratio measurement here is based on the 50 Ω impedance of R1 receiver in the instrument, and the direct response calibration must be carried out at TP1 test point in order to correctly measure the voltage transfer function / out / / in. If the through response calibration is not performed (or the feedthrough is not connected, as shown in Figure 5), the measured gain will be 6 dB higher than the correct value because the AC voltage measured by the internal 50 Ω reference receiver is only half of VIN.

When measuring in a high frequency range of more than tens of MHz, connecting a 50 Ω feedthrough to the input port of the device under test can prevent the standing wave caused by the impedance mismatch between the 50 Ω impedance of the instrument and the high input impedance of the device under test. However, connecting the feedthrough will form a shunt signal path between the central conductor and the grounding of the measuring cable, which may produce measurement errors related to the grounding loop when measuring large Attenuators (such as CMRR and PSRR), so attention must be paid. If strictly considered, it is best not to connect feedthrough.

Figure 5 configuration of measuring amplifier using S-parameter test port and active probe (maximum measurement frequency up to 30 MHz)

Non 50 Ω DUT, example 2

Figs. 6 and 7 are configuration examples for measuring a 2-port device, and the input and output impedance of the device ranges from hundreds of Ω to 1 or 2 K Ω. Typical applications are low frequency passive filters, such as ceramic filters and LC filters. In these examples, impedance matching can be achieved by connecting only one series resistor. Figure 6 shows the configuration of the test using the gain phase test port. The ratio VT / VR is the transmission coefficient of the 1 K Ω system impedance.

In the measurement of some filters, it is necessary to connect a load capacitor CL in parallel with the load resistance before testing. In order to prevent the influence on the characteristic parameters of the filter in the measurement, the input capacitance of the high impedance probe must be very low. Therefore, the high impedance t-channel receiver should be connected with a 10:1 passive probe with an input capacitance of about 10 PF. Otherwise, if the device under test is sensitive to capacitive load, it should be measured on the S parameter test port of the instrument with an active probe. Please see the configuration of the measurement amplifier shown in Figure 5.

Equivalent measurement results can be obtained by using the 50 Ω internal resistance of the t-channel instead of the high impedance probe, and connecting another matching resistor as shown in Fig. 7. This configuration is simpler and has the advantage that the capacitance of the probe will not be introduced into the T channel. However, this configuration is not suitable for measuring filters with high rejection ratio, because the series matching resistance will reduce the dynamic range of the measurement. In this case, the dynamic range will decrease by 20 * log (50 / 1000) = 26 dB.

Fig. 6 measurement configuration of passive if filter with high impedance probe (when the tested device is not very sensitive to capacitive load)

Figure 7 measurement configuration of passive if filter using instrument 50 Ω input port

Use the probe to measure directly on the circuit board

The second application example is to use the probe to measure directly on the circuit board. First, measure the frequency response characteristics of the circuit or device between two test points on the tested circuit board. Fig. 8 shows how to measure the frequency response characteristics of the circuit module 2 using the gain phase test port. By using two high impedance probes to detect at TP1 and TP2 test points, the frequency response characteristics of circuit module 2 can be directly measured.

Similar to the configuration of the measuring amplifier in Figure 4, when connecting the high impedance receiver of the instrument with the tested device, the coaxial test cable or 10:1 passive probe shall be properly selected according to the maximum test frequency, the input impedance of the probe and the input capacitance of the probe.

Figure 8 measurement of the device under test on the circuit board using the gain phase test port and two high impedance probes (maximum test frequency up to 30 MHz)

The maximum test frequency of e5061b vector network analyzer} gain phase test port is 30 MHz. If the frequency of using the probe to measure the devices on the circuit board exceeds 30 MHz, the solution is to connect an active probe to the S parameter test port, and then complete the measurement in two steps as shown in Figure 9.

Firstly, the active probe point is on the TP1 measuring point to measure the response characteristics of circuit module 1, and the measurement results are stored as register tracks. Then measure the overall response characteristics of circuit modules 1 and 2 on the TP2 measuring point, and store the measurement results as data tracks. Finally, we can use the instrument to calculate the data track / register track to obtain the frequency response characteristics of circuit module 2.

If the probe point is calibrated on the TP1 measuring point first, and then the probe point is measured on the TP2 measuring point, equivalent measurement results may also be obtained. In this way, the response characteristics of circuit module 2 relative to TP1 reference point can be obtained directly without using the operation function of trajectory.

If the output characteristics of the device under test at TP2 point are sensitive to the capacitance of TP1 point, the conditions of the device under test in the second step will be slightly different from the first step, and there will be errors in the final measurement results obtained from the calculation of the measurement results of these two steps. In order to minimize the measurement error, as shown in Fig. 9, only in the second step of measurement, it is necessary to connect a virtual capacitance C2 whose capacitance value is roughly equivalent to the input capacitance of the active probe. One of the applications of this capacitance compensation method is to use the above two-step measurement method to measure the phase margin of high-speed operational amplifier. We will introduce an actual measurement example later.

Figure 9 measuring devices in a circuit board using a high impedance probe (maximum test frequency up to 30 MHz)

If bandwidth (ifbw) setting for low frequency measurement

How to set ifbw (intermediate frequency bandwidth) in measurement is one of the common problems encountered by many users of low frequency network analyzer. A wider ifbw is generally used for high-frequency measurement to obtain faster scanning speed, but a narrower ifbw is required for low-frequency measurement to avoid measurement errors mainly caused by lo feedthrough. Taking a device with large attenuation as an example, assuming that the starting frequency of the measurement is 1 kHz and the ifbw is 3 kHz, the small signal attenuated by the device under test will be up converted to an intermediate frequency (if) signal and can pass through the IF filter of the receiver. At this time, there will be a problem. As shown in Figure 10, the frequency of the leakage signal (LO feedthrough) of the local oscillator is also very close to the IF frequency, and it can also pass through the IF filter, which will cause untrue frequency response measurement results.

Figure 11 shows the measurement results of a 60 dB attenuator measured with the gain phase test port of e5061b. The power of the measurement signal is - 10dBm, the measurement starting frequency is 1kHz, ifbw is set to 3kHz, and the attenuators of T measurement channel and R measurement channel are set to 20dB. You can see from the measurement results displayed that there is an erroneous measurement response caused by lo feedthrough near the starting frequency. Even when measuring devices such as low-pass filters and the measured RF signal power is high, a similar situation will occur.

In this case, the trajectory measured near the starting frequency will become unstable due to the interference of Lo feedthrough which is very close to the RF signal frequency. To avoid these problems, you can set the ifbw to a value much lower than the starting frequency (for example, set it to 1 / 5 of the starting frequency), or use the mode I of ifbw auto (if bandwidth automatic). When the instrument performs logarithmic scanning, the value of ifbw will be automatically set from narrow to wide every ten times of frequency change, so that the total scanning time will not be too long. The ifbw auto mode of e5061b sets the value of each ifbw to one fifth of the starting frequency of every ten octave band with the increase of scanning frequency.

Figure 10 measurement error caused by lo feedthrough

Fig. 11 measurement results of 60 dB attenuator (start frequency = 1 kHz, ifbw = 3 kHz and auto)

Measurement method using high impedance probe

Proper detection method is very important for accurate measurement with high resistance probe. Special attention should be paid to the input capacitance of the probe. The large input capacitance on the probe will reduce the input impedance of the probe under high-frequency measurement conditions. For example, if the input capacitance (CIN) of the probe tip is 100pF, its input impedance is 15.9 K Ω (1 / (2 * pi * f * CIN)) when the measurement frequency is 100kHz, which is still high impedance. However, if the measurement frequency rises to 10 MHz, its input impedance becomes 159 Ω. For many measurement cases, this impedance is not high enough. In addition, the high input capacitance of the probe will also affect the measurement results of devices sensitive to capacitive load, such as passive if filter, resonant circuit and some parameters of the amplifier determined by capacitance conditions (such as the phase margin of the amplifier). For these applications, if the network analyzer has a high impedance input port (e.g. e5061b), it is necessary to use the detection method of low input capacitance. The simplest way to connect the DUT during measurement is to connect the DUT to the high impedance input port of the instrument using a coaxial cable (e.g. BNC cable with a test clamp at one end) or a 1:1 passive probe, as shown in Figure 12.

If the measurement frequency range is lower than 1 MHz, and the input capacitance of the probe as a capacitive load will not affect the device under test, this method is a good solution. Compared with the 10:1 passive probe, this 1:1 detection method will not reduce the dynamic range of measurement, and can have a good signal-to-noise ratio (SNR) even for small signals. The disadvantage of this method is that the input capacitance of the probe will be high due to the superposition of the test cable capacitance and the capacitance of the high impedance input port. Even if a very short cable is used, the input capacitance at the end of the cable will reach dozens of PF. Therefore, this method is not suitable for high-frequency measurement with frequency more than 1MHz, nor for measurement sensitive to capacitive load.

Figure 12 coaxial test cable or 1:1 passive probe

As shown in Figure 13, the 10:1 passive probe commonly used in oscilloscope can reduce the probe input capacitance. This probe is specially designed for use with high impedance input port. 10: 1 the input capacitance at the end of the passive probe is generally about 10PF, which enables it to be used for higher measurement frequency detection. Similar to the application of general oscilloscope, if there is a high input impedance measurement port in the instrument, it is a common way to use 10:1 passive probe for high impedance detection. Its disadvantage is that the measurement dynamic range will be reduced by 20dB due to the influence of probe 10:1 attenuation. Therefore, this method is not suitable for measuring very small signals.

The active probe has high input resistance and very small input capacitance, and because there are active circuit components near the port of the probe, it will not attenuate the measured signal, as shown in Figure 14. For example, the input resistance / / capacitance of 41800a active probe (from DC to 50 Ω MHz) is 100 K Ω / / 3pf respectively. In addition, you can connect a 10:1 adapter at the end of the probe to make the impedance and capacitance of the probe reach 1 m Ω / / 1.5 PF, but this will reduce the dynamic range by 20 dB. If you need to measure in a high frequency range of more than 30 MHz, or the device under test is very sensitive to capacitive load, we recommend you choose an active probe.

Figure 13 10:1 passive probe

Figure 14 active probe

Separation of signals in ratio measurement

In order to measure the transmission coefficient of 50 Ω devices, such as the passive filter with system impedance Z0 = 50 Ω, or the transmission coefficient of devices with characteristic impedance Z0 of other values (the system impedance needs to be converted by matching circuit), it is necessary to separate the signals output by the excitation source of the instrument and send them to the r-channel measurement receiver (reference signal) of the instrument 50 Ω and the input port of the tested device respectively. If the output port of the excitation source used does not have a built-in signal separation device (e.g. built-in power separator or built-in directional bridge), it is necessary to use an appropriate separation device to complete the signal separation outside the instrument.

E5061b-3l5 has S-parameter test port. For the measurement of transmission characteristics of most 50 Ω devices, S-parameter measurement port can be used without external signal separation devices. However, in some applications that need to use the gain phase test port of the instrument to measure the transmission coefficient, such as measuring the output impedance of DC-DC converter by shunt thru method, it is necessary to use external signal separation devices.

For general network analysis based on measuring linear devices, the most important requirement for signal separation devices is to ensure 50 Ω excitation source output impedance (source matching) during ratio measurement. The most common and recommended signal separation device is the dual resistance power separator, with a frequency range from DC to GHz, which can ensure excellent source output impedance in ratio measurement.

The ratio measurement using the power separator shown in figure 15-a is equivalent to the two measurements completed in figure 15-b. the AC voltage (VO) at the branch point on figure 15-a can be regarded as the two virtual excitation source voltages on figure 15-b. As shown in the figure, the equivalent source output impedance in r-channel and t-channel measurement is 50 Ω, which is usually the ideal source matching condition for 50 Ω network measurement.

Please note that the dual resistance power separator is only suitable for ratio measurement, not for absolute voltage measurement of 50 Ω system impedance, because the physical output impedance of the separator is 83.3 Ω from the direction of the tested device, not 50 Ω.

FIG. 15 line ratio measurement of 50 Ω device using power separator

In addition to the power separator, other devices that can separate signals are low-frequency directional coupler or reactive power distributor (AC coupling with transformer), and their two output ports have high isolation (25 or 30dB). Zfdc-15-6 directional coupler (0.03 to 35 MHz, BNC interface) or ZFSC power distributor (0.002 to 60 MHz, BNC interface) produced by (minicircuits. Com) is one of the representative products. Although their maximum frequency is only about 30 MHz or 60 MHz, and the low-frequency frequency can only reach a few kHz or tens of kHz, these devices are ideal choices when the frequency range can meet the application requirements. Because of the high isolation between their two output ports, the reflected signal from the input port of the tested part will not directly enter the r-channel receiver, so it will not affect the measurement results of the r-channel.

If the above devices are used as signal separation devices in ratio measurement, the effect of their equivalent source matching will not be as good as that of using double resistance power separator. In order to improve the effect of source matching, it is sometimes necessary to connect an attenuator (about 6 dB) between its output port and the device under test. The advantage of this signal separation device over the power separator is that its absolute source output impedance (Port matching) is 50 Ω, which enables you to measure the absolute voltage in a 50 Ω environment, although in general, the absolute voltage measurement in low-frequency measurement applications is not as meaningful as that in RF applications.

The resistance value of the three resistance arms of the resistance power distributor composed of three resistors is Z0 / 3. This power separator is not suitable for ratio measurement. If we take the branch point of the three resistance power divider as the virtual signal source (similar to the double resistance power divider), the equivalent source output impedance is not 50 Ω, but 50 / 3 = 16.7 Ω, and the isolation between the output ports is low (only 6dB). Unless the input impedance of the device under test is accurate to 50 Ω, the use of three resistance power divider in ratio measurement will produce serious measurement error.

Figure 16 directional coupler / Bridge

Figure 17 resistance power divider (not applicable to ratio measurement)

Measurement of large attenuation devices in low frequency range

Measurement error

For devices with large attenuation measured by traditional low-frequency network analyzer, when the measurement frequency is below 100 kHz, the measurement results are likely to be affected by the errors related to the grounding loop of the test cable. These errors will be obvious when measuring CMRR and PSRR of low frequency amplifier. The most serious problem is the error caused by measuring the shielding resistance of the cable (the resistance of the metal braid), which can not be ignored in the low frequency range below 100 kHz.

Fig. 18 is a case of measuring a large attenuation device using a network analyzer. When the attenuation value of the device under test is very high, the output voltage Vo of the device under test will be very small. Ideally, the AC voltage measured by the measuring receiver VT should also be vo.

However, in the low frequency range, external common mode noise is likely to enter the grounding loop of the test cable between the excitation source and the receiver, as shown in Figure 18. The voltage drop on the resistance RC2 of the outer shielding layer of the measuring cable is vc2. Since the measured voltage Vo itself is a small value, the voltage vc2 will cause the voltage measurement error of the receiver VT, so the final measured attenuation value will be wrong.

According to the different phase relationship between VO and vc2, the actual measured attenuation value may be higher or lower than the real attenuation value of the device under test. Or in some cases, there will be an obvious subsidence on the trajectory of the measurement results.

Fig. 18 measurement error caused by cable shield resistance (1)

The grounding loop of the test cable will cause additional measurement errors in the low-frequency measurement range. You can imagine that the device under test has a shunt signal path and its impedance Zsh is very small. A typical example is to use the shunt thru method to measure the milliohm impedance of components on the power distribution network in the low frequency band, such as the impedance of DC-DC converter and large capacity bypass capacitor.

Ideally, the signal of the excitation source should be returned to the excitation source side through the external shielding metal of the measuring cable after passing through the device under test.

However, during the low-frequency test, the current of the excitation source will also flow into the shielding layer of the test cable on the side of the t-channel measurement receiver. Similar to the phenomenon of common mode noise, the excitation source current flowing into the shielding layer of the t-channel measuring cable will produce a voltage drop vc2 on the resistance RC2 of the outer shielding layer of the measuring cable, which will cause errors in the measurement results of the receiver vt. In this case, the measured attenuation value will be greater than the real attenuation value of the tested part.

It should be noted that these measurement errors related to the grounding loop of the test cable will only occur in the range of measurement frequency lower than 100 kHz. In the higher measurement frequency range, the inductance of the coaxial test cable acts as a common mode choke (balun), so that the current causing the measurement result error will not pass through the shielding layer of the measurement cable on the side of the VT receiver.

Fig. 19 measurement error caused by shield resistance of cable (2)

Measurement of large attenuation devices in low frequency range

Traditional solutions

At present, there are several techniques to minimize the measurement error described above. Traditionally, the most commonly used method is to sleeve the small magnetic ring on the test cable or wind the test cable on the large magnetic ring for several turns. The equivalent circuit using the magnetic ring method is shown in Figure 20. The magnetic ring can increase the impedance of the measuring cable shield and suppress the current flowing through the cable shield without affecting the current flowing into the central conductor of the measuring cable and returning to the side of the excitation source.

The impedance generated by the inductance of the magnetic ring itself on the shielding layer of the measuring cable will reduce the common mode noise current flowing through the grounding loop and the excitation source current flowing into the shielding layer of the measuring cable on the side of the VT receiver. In addition, a magnetic ring is also used on the measuring cable on the side of the excitation source to return the excitation source current to the side of the excitation source through the shielding layer of the cable.

But in fact, this method is not easy to do, because we need to find a high-quality magnetic ring with high inductance (high permeability), so that it can completely eliminate the error in the very low measurement frequency range. In addition, it is sometimes difficult to judge whether the magnetic ring is working effectively, especially when the attenuation characteristics of the device under test are uneven.

For this application, the ring core we recommend is metglas Finemet f7555g（ Φ  79 mm) 。 Please refer to www.metglas.com com.

FIG. 20 solution of using magnetic ring to reduce measurement error

Solution using e5061b-3l5

The gain phase test port (5 Hz to 30 MHz) of e5061b-3l5 has a unique hardware architecture, which can eliminate the measurement error caused by the grounding loop of the test cable from the signal source to the receiver. Fig. 21 is a simplified block diagram of measurement using a gain phase test port. The receiver is connected in series with a semi floating impedance | ZG | which is about 30 Ω in the low frequency range below 100 kHz. Similar to the case of using a magnetic ring, we can intuitively see that the impedance | ZG | prevents the shielding current of the measuring cable. Alternatively, we assume that the voltage swing on the grounding side of the device under test is VA, as shown in Figure 21. Since rshieid is much smaller than the input impedance of the receiver by 50 Ω, VT can be approximately obtained by the following formula:

VT=Vc2+Vo=Va x Rc2/(Rc2+Zg）+Vo

Because RC2 < < ZG |, the first term in the above formula can be ignored, VT is almost the VO we really need to measure. Therefore, by minimizing the influence of shielding resistance, the large attenuation or milliohm parallel impedance of the device under test can be measured correctly. The gain phase test port of e5061b can easily and accurately measure large attenuation values in low frequency range.

On the other hand, like other existing low-frequency network analyzers, the measurement receiver of S-parameter test port of e5061b-3l5 adopts standard grounding architecture. If the S-parameter test port (for example, when the measurement frequency exceeds 30 MHz and the gain phase test port cannot be used for measurement) is used to measure the low-frequency large attenuation device, the magnetic ring still needs to be used to eliminate the error caused by the grounding loop of the test cable.

Figure 21 solution using e5061b - 3l5 gain phase test port

Effectiveness of gain phase test port

Fig. 22 shows the transmission measurement results of 90 dB coaxial attenuator with e5061b ＾ s parameter test port and gain phase test port. The test frequency range is 100 Hz to 10 MHz. The measurement track of channel 1 on the left in the figure is the measurement result of the S-parameter test port. As shown in the figure, the measurement results without using the magnetic core show incorrect measurement results with large values in the low frequency band, which is caused by the error caused by the grounding loop of the test cable between the excitation source and the receiver. Another track in the same figure is the measurement result after adding a magnetic ring to the test cable. Although the measurement result in the low frequency band is improved, the measurement result in the very low frequency band is still not accurate enough.

The measurement track of channel 2 on the right side of the figure is the measurement result using the gain phase test port. As shown in the figure, this method can correctly measure the attenuation of - 90dB when the measurement frequency is below 100 Hz, and the measurement results will not be affected by the grounding loop of the test cable.

Fig. 22 comparison of measurement results obtained by three different measurement methods

Operational amplifier measurement example

Closed loop gain

The following sections detail examples of measuring various frequency response characteristics of operational amplifiers.

Figure 23 shows an example of measuring the closed-loop gain configuration of a simple inverting amplifier (AV = - 1) with a gain phase test port (measurement frequency up to 30 MHz).

In order to minimize the influence of probe capacitance on amplifier load conditions, it is recommended to use 10:1 probe, which has relatively small input capacitance.

In order to accurately measure the frequency response characteristics of gain and phase, it is necessary to calibrate the probe point of T measurement channel on TP1 test point, so as to eliminate the gain and phase errors between the two probes.

Fig. 23 configuration example of ring gain measurement using gain phase test port

If you need to measure the frequency response characteristics of the amplifier at frequencies above 30 MHz, you need to use the S-parameter test port and active probe. Figure 24 shows a configuration example. We must calibrate the probe point on the TP1 test point for direct response. Because the input impedance of the receiver of channel R is 50 Ω, we need to set the reference point on TP1 so that we can measure the voltage transfer function of the input and output ports of the device under test.

Figure 25 shows an example of closed-loop gain measurement of high-speed operational amplifier with S-parameter test port of e5061b and 41800a active probe. The cursor is located at the cut-off frequency of - 3 dB, which indicates that the bandwidth of the amplifier circuit is about 20 MHz.

Fig. 24 configuration example of ring gain measurement using S-parameter test port

Frequency = 100Hz to 100MHz
Excitation source power = 0dbm
If bandwidth automatic (upper limit = 1kHz)

Figure 25 example of closed loop gain measurement

Open loop gain

There are many methods to measure the open-loop gain of operational amplifier. The most common method is to measure the voltage ratio VT / VR in the circuit, as shown in Figure 26. Assuming that the open-loop gain of the operational amplifier is a, if the current is IR2, the following formula can be obtained:

(VT-VR)/R2 = {VT-(-A x VR)}/Zout

If zout < R2, the voltage ratio VT / VR can be calculated according to the following formula

VT/VR = (-A-Zout/R2)/(1-(Zout/R2)) = -A

For high gain operational amplifiers, if the closed-loop gain AV is very small (e.g. AV = - R2 / R1 = - 1), the voltage VR will be too small to be measured accurately, especially when the open-loop gain is very high in the low frequency range.

In the linear working area, if the closed-loop gain AV increases, the voltage VR will also increase proportionally, and it will be easier to measure with an instrument. For example, if AV = R2 / R1 = 10, VR will be the value of VR when AV = 1.

Fig. 26 configuration example of closed loop gain measurement

Fig. 27 shows a configuration method for measuring with a gain phase port. The result of ratio measurement T / R can directly represent the open-loop gain a. In order to accurately measure the frequency response characteristics of the phase without being affected by the load conditions caused by large probe capacitance, a 10:1 passive probe should be used instead of a coaxial test cable.

Fig. 27 configuration example of open loop gain measurement using gain phase test port

Fig. 28 shows the measurement results of the open-loop gain of the operational amplifier under the unit gain condition (R1 = R2 = 1 K Ω) measured by the gain phase configuration method of Fig. 27, and the test frequency range is from 10 Hz to 30 MHz. The phase margin can be derived from these measurements. Assuming that there is no phase shift, simply find the feedback path transfer function 阝: RI / () I + R2) = & frac12; =- 6 dB line, and then place a cursor on the + 6 dB point to find the loop gain I-A & times; 阝] = 0 dB intersection. The phase margin can be directly read out from the corresponding position of the cursor on the phase track, just like the cyclic transfer function ax 阝 (including 180 degree inversion) we see at the input port of the operational amplifier.

The trace fluctuation in the high gain region is caused by the dynamic performance degradation caused by the 20 dB loss of the passive probe. Because we measure the open-loop gain under the unit gain of the amplifier, the AC voltage measured by the r-channel receiver will be very small in the high gain region, which will lead to the fluctuation of the trace. The trace fluctuation in the high gain region is not a problem for measuring the phase margin of the measured data in the low gain region.

However, if you also want to measure very high gain in the low frequency range, you need to replace the 10:1 passive probe with a coaxial test cable and measure another open-loop gain separately. The attenuator of the R-port receiver shall be set to 0 dB and the attenuator of the T-port receiver shall be set to 20 dB, so that a very small voltage can be measured on the r-channel receiver under the condition of very good signal-to-noise ratio. Please note that this measurement configuration is only applicable to the medium and low frequency range, where the open-loop gain is relatively high, and the voltage on the r-channel receiver will not exceed the maximum input level of the receiver (the attenuator is set to 0d).

Fig. 28 example of open loop gain and phase measurement using gain phase ports

If the open-loop gain of the operational amplifier is measured at more than 30 MHz, the active probe and S-parameter test port shall be used. Since only one active probe is allowed for the S-parameter test port, you need to use the two-step measurement method. The specific steps are as follows:

1．Calibrate the response of the probe on the TPI test point.

2．Measure S21 with the probe point on the TP2 test point, and store the track line data through data - > MEM operation (the first step of measurement).

3.Connect a virtual capacitor to TP2, and then measure S21 at TP3 test point (the second step of measurement).

4. Using the mathematical function calculation function of the instrument, divide the measurement result of the second step by the data (data / memory) already stored in the register of the first step to obtain the result of open-loop gain.

The virtual capacitance connected in the second step measurement is the same as the probe capacitance in the first step measurement. Within the high frequency measurement range, it will affect the measurement structure of the open-loop phase. The capacitance of this virtual capacitance should be the same as the input capacitance of the active probe.

If you need to measure a very high open-loop gain, it is best to use a magnetic ring on the test cable to eliminate the measurement error caused by the grounding loop, which may affect the measurement results of very small signals in the first step.

Fig. 29 configuration example of open loop gain measurement using an active probe

Fig. 30 shows an example of measuring open circuit gain and phase with the configuration in Fig. 29. Track 1 is the response result measured at the TP2 test point. It is the ratio of the input voltage to the attenuated voltage at TP2. Track 2 is the response measured at the TPB test point, which is the closed-loop gain and phase. Trajectory 3 is the open-loop gain and phase calculated from these measurement results, which are obtained by performing mathematical function calculation (data / memory) on the measured trajectory.

As mentioned earlier, the phase margin is the value of the corresponding phase measurement result when the open-loop gain is equal to 6 dB. At this time, the loop gain is 0 dB. In this example, the phase margin is about 86 degrees.

Figure 30 example of open loop gain and phase measurement using an active probe

Common mode rejection ratio CMRR

CMRR (common mode rejection ratio) of operational amplifiers and other low frequency differential amplifiers is usually difficult to measure because you need to measure very large common mode input attenuation. The common mode rejection ratio is defined as CMRR = ad / AC, where ad is the differential mode gain and AC is the common mode gain. Figure 31 shows the configuration of measurement with gain phase test port. In order to measure large attenuation values, it is necessary to connect the receiver and the tested part with a coaxial test cable instead of a 10:1 passive probe with 20 dB loss.

You can turn switch SW1 to position a to measure common mode gain (attenuation) AC and SW1 to position B to measure differential gain ad. Then, CMRR is calculated according to AD / AC (= 20 & times; log (AD / AC) in DB). The differential gain of the circuit is IADI = R2 / R1 = 10, and its common mode gain AC is 10 times (i.e. 20 dB) when IADI = 1. This measurement method can make the instrument measure CMRR of more than 100 dB.

Because the gain phase test port is a semi floating receiver architecture, you can accurately measure the high CMRR by eliminating the measurement error caused by the test cable grounding loop.

Fig. 31 configuration example of common mode rejection ratio CMRR measurement using gain phase test port

CMRR with frequency higher than 30 MHz can be measured using S-parameter test port and active probe. In this case, it is necessary to use a magnetic ring on the test cable, as shown in Figure 32, to eliminate the measurement error caused by common mode noise. Metglas Finemet f7555g magnetic ring can be used（ Φ  79 mm： metglas. com )。

Figure 33. Shows an example of measurement with a gain phase test port. Track 1 represents common mode gain AC and track 2 is differential mode gain ad (= 20dB). By eliminating the influence of the grounding loop, the common mode gain AC of about - 90 dB can be accurately measured. Trajectory 3 is the CMRR calculated from these results. The cursor on it indicates that the CMRR is about 80 dB at 100 kHz. In the low frequency range, CMRR is greater than 100 dB.

Figure 32 configuration example of CMRR measurement using S parameter port

Frequency = 100Hz to 100MHz
Excitation source power
For AC measurement: 0 dBm
For ad measurement: - 15 dBm
If bandwidth = automatic (max. 100 Hz)
Receiver att settings
AC measurement: 20 dB (R channel)
0 dB (t-channel)
Ad measurement: 20 dB (r-channel and t-channel)
In this measurement example, the balance between RI and R2 is not fully optimized.

Figure 33. Example of CMRR measurement using gain phase port

Power rejection ratio (PSRR)

Power rejection ratio (PSRR) of amplifier is another difficult parameter to measure because it requires measurement of large attenuation value. Here, it is defined as PSRR = AV / AP, where AV is the closed-loop gain of the amplifier circuit and AP is the gain from the input port (positive / negative) of the power supply to the output port. Similar to CMRR measurement, AP is directly proportional to AV in the linear operating range.

Fig. 34 shows a configuration example of measuring PSRR (positive PSRR) with a gain phase port. Since IAVI = R2 / R1 = 1, the measured circuit gain is directly indicated as the reciprocal of PSRR (= 1 / AP, a DB value with a negative value) of the operational amplifier. The measured excitation source signal is applied to the positive pole of the power supply and has a DC bias voltage. The e5061b has a built-in DC bias source that allows you to internally attach a DC voltage bias to the AC signal of the excitation source.

Fig. 34 configuration example of PSRR measurement using gain phase test port

PSRR with frequency higher than 30 MHz can be measured using S-parameter test port and active probe. Similar to the CMRR measurement using the S-parameter test port, we recommend using the magnetic ring on the test cable to eliminate the measurement error caused by the grounding loop of the test cable. Figure 36 shows an example of PSRR measurement with a gain phase test port. The cursor on it shows that the PSRR is about - 87 dB at 1 kHz. E5061b-3l5 has DC monitoring function, which can be used to check the value of DC voltage actually applied to the tested device.

Figure 35 configuration example of PSRR test using S parameter test port

Fig. 36 example of PSRR measurement using gain phase test port

Output impedance

The measurement of output impedance of operational amplifier is not the measurement of two port transmission parameters, but the measurement of single port impedance. Typically, the closed-loop output impedance of operational amplifiers ranges from tens of milliohms at low frequencies to 100 ohms at high frequencies. In order to measure completely within this impedance range, reflection measurement method will be an appropriate solution. Fig. 37 shows a configuration example for measuring the closed-loop output impedance of an operational amplifier. Open circuit / short circuit / load calibration (full single port calibration) must be done during measurement.

Fig. 37 configuration example of output impedance measurement

Fig. 38 is a measurement example of closed-loop output impedance. The measured trace shows the amplitude of the impedance value plotted through the calculation of the impedance conversion function. The trace on the left shows the output impedance in logarithmic scale [20x log izi DB]. The trace on the right shows the output impedance in linear scale [Ω].

Figure 38 example of output impedance measurement

This content comes from the network / high-speed RF Baihuatan. This website only provides reprint. The views, positions and technologies of this article have nothing to do with this website. If there is infringement, please contact us to delete it!